Multiple channels of data can share a single transmission medium, but a receiver may desire data from one channel. Therefore, to isolate the desired channel for processing, transceivers typically perform at least three operations on the received signal: (1) the undesired channels are filtered out; (2) the desired channel is “shifted” to dc, where it can be processed; and (3) the signal is amplified. The order of the operations depends on the design of the transceiver. Shifting a signal may be accomplished by mixing the signal with a local oscillator signal.
In superheterodyne transceivers, the input signal (e.g., a Radio Frequency (“RF”) signal) is amplified and filtered. Then, the filtered RF signal is shifted to an intermediate frequency (“IF”) where it is passed through a highly selective filter and substantially amplified before it is shifted to dc for processing.
Direct conversion transceivers use techniques to avoid having to use an IF, thereby saving power, cost and allowing for a smaller physical design for some applications (e.g., mobile communications devices). A part of an exemplary direct conversion (zero-IF) receiver 100 is illustrated in FIG. 1. The receiver 100 includes an antenna (A) 102, a low-noise amplifier (LNA) 104, two mixers 106, 108, two real low-pass filters (LPFs) 110, 112, two analog-to-digital converters (ADCs) 114, 116, and a Fast Fourier Transform (FFT) module 118. The FFT module 118 may be included, e.g., as part of an exemplary Orthogonal Frequency Division Multiplexer (OFDM) Digital Signal Processor (DSP), used in the application of 802.11a, 802.11g or another OFDM standard.
Antenna 102 receives signal, xIN, which is fed to the input of LNA 104. LNA 104 amplifies signal xIN and outputs signal xA. Signal xA is input in parallel to each of the mixers 106, 108. Mixer 106 mixes signal xA with local oscillator signal, lo1 (“I”) and outputs signal x1. Mixer 108 mixes signal xA with local oscillator signal, lo2 (“Q”) and outputs signal x2. The two local oscillator signals, lo1 (“I”) and lo2 (“Q”) are quadrature related (separated in phase by 90°). Signal x1 is filtered by low-pass filter 110, outputting filtered signal y1. Signal x2 is filtered by low-pass filter 112, outputting filtered signal y2. Filtered signals y1, y2, are input to ADCs 114, 116, respectively, which sample the analog filtered signals y1, y2 at a sampling frequency, fs. The digital outputs from ADCs 114, 116 are input to FFT 118 which processes and reforms the transmitted signal in the digital domain.
An exemplary zero-IF transmitter (TX) chain 200 is shown in FIG. 2. Transmitter (TX) chain 200 starts with a DSP 202, followed by a pair of digital-to-analog converters (DACs) 204, 206, a pair of LPFs 208, 210, a mixer/LO block 212, a power amplifier (PA) 214, and an antenna 216. DSP 202 generates and outputs pairs of digital quadrature signals, which are input to the pair of DACs 204, 206. The DACs 204, 206, operating at a sample rate fs, convert the digital quadrature signals to analog quadrature signals. The analog quadrature signals are input to the pair of real low pass filters 208, 210. The real low pass filter pair 208, 210 filter the analog signals, and the filtered analog signals are input to the mixer/local oscillator block 212. Mixer/lo block 212 mixes LPF 208 output signal with lo1 (I), mixes LPF 210 output signal with lo2 (Q), and combines the two mixed signals. The output from mixer/lo block 212 is fed to a power amplifier 214, which amplifies the signal prior to transmission over antenna 216.
The two quadrature paths (I path and Q path), corresponding to the I and Q local oscillator signals, allow the direct conversion receiver/transmitter to avoid having to use an IF. Unfortunately, the characteristics of direct conversion receivers, transmitters, and transceivers are not ideal.
I/Q mismatch (sometimes referred to as I/Q errors) causes I/Q imbalance (sometimes referred to as I/Q leakage). I/Q imbalance in receivers, transmitters, and transceivers, an area of concern to the present invention, shall now be described. Main contributors of the RX 100 chain's (FIG. 1) I/Q imbalance are the gain error (δG1) 105 of the mixers (106, 108), the phase error (δP1) 107 in the local oscillator signals, the gain (δG2(ω)) 109 and phase (δP2(ω)) 111 mismatch between the LPF's (110,112) transfer functions, and the gain error (δG3) 115 between the ADCs (114,116).
Slight differences in the components of the mixers 106, 108 may contribute to the gain error (δG1) 105. Slight differences in the I/Q relationship of the lo1 and lo2, and the I and Q signals that are not exactly in quadrature may cause phase error (δP1) 107. Component mismatches between filters 110, 112 may cause the gain (δG2(ω)) 109 and phase (δP2(ω)) 111 errors. If the components of filters 110, 112 are not perfectly matched, (i.e., if the transfer functions do not match (H1(ω)≠H2(ω), where H1(ω) represents the transfer function for LPF 110 and H2(ω), represents the transfer function for LPF 112)) then a non-zero transfer function Hdf(ω), contributes to a leaked (undesired or difference) output component. Even when the filters are fabricated at the same time and on the same integrated chip, component mismatch of 0.2% to 0.5% or even larger may still occur. A parallel model of an imperfect low-pass filtering operation is illustrated in FIG. 3. The top branch represents the common component, hcm, of H1(ω) and H2(ω), which produces the desired output. The bottom branch represents the difference component, hdf, between H1(ω) and H2(ω), which produces the leaked signal. Returning to FIG. 1, slight differences in the components of the ADCs 114, 116 may contribute to the gain error (δG3) 115.
The I/Q imbalance contribution of gain and phase errors can be modeled as a two-input two-output linear network with some inter-coupling coefficients. These simple models can be individually applied to each block of mixers/lo, LPFs and ADCs, as shown in error model representations 120, 122, and 124, respectively of FIG. 1. Note that the LPF error model representation coefficients are a function of frequency, which significantly complicates the modeling and compensation.
Main contributors of the TX 200 chain's (FIG. 2) I/Q imbalance are the gain error (δG1) 211 of the mixer/lo block 212, the phase error (δP1) 213 in the local oscillator signals, the gain (δG2(ω)) 207 and phase (δP2(ω)) 209 mismatch between the LPF's (208, 210) transfer functions, and the gain error (δG3) 205 between the digital-to-analog data converters (204, 206). The I/Q imbalance contribution of gain and phase errors can be modeled as a two-input two-output linear network with some inter-coupling coefficients. These simple models can be individually applied to each block of mixers/lo, LPFs and DACs, as shown in error model representations 218, 220, and 222, respectively of FIG. 2. Note that the LPF error model representation coefficients are a function of frequency, which significantly complicates the modeling and compensation.
FIG. 4 illustrates the concepts of leakage and compensation which may be applied to I/Q receivers, transmitters and transceivers using two-input two-output leakage and correction models.
In mathematical terms a leakage stage 404 may be described as:{right arrow over (y)}=Aleak×{right arrow over (x)}  (1)where {right arrow over (y)}=[y1,y2]T (output), {right arrow over (x)}=[x1,x2]T (input) and Aleak (model) are given in FIG. 4. The resulting image rejection ratio (IMR) corresponding to the leakage stage 404 can be calculated by:
  IMR  ≅      10    ⁢                  ⁢          log      10        ⁢                                                (                          1              +                              δ                ⁢                                                                  ⁢                                  G                  2                                            +                              2                ⁢                δ                ⁢                                                                  ⁢                G                ⁢                                                                  ⁢                cos                ⁢                                                                  ⁢                δ                ⁢                                                                  ⁢                P                                      )                                (                          1              +                              δ                ⁢                                                                  ⁢                                  G                  2                                            +                              2                ⁢                δ                ⁢                                                                  ⁢                G                ⁢                                                                  ⁢                cos                ⁢                                                                  ⁢                δ                ⁢                                                                  ⁢                P                                      )                                      ⁡              [        dB        ]            
The concept of IMR shall now be described. An imperfect quadrature signal can be modeled in the phase domain as two rotating phasors with ω0 angular frequency by (n/2−φ) apart, and with AI and AQ magnitudes. The frequency domain representation of this two-path signal contains a desired component at ω0 and a leakage (undesired) component at −ω0. The image rejection ratio (IMR) may be used as a representation for the degree of I/Q imbalance. The IMR would be infinitely large (desirable) if the gain imbalance γ=AI/AQ was unity and the phase imbalance φ was zero; such an IMR (corresponding to leakage stage 404) would be represented in the Aleak model representation with δG=1 and δP=0.
The concept of I/Q imbalance compensation is straightforward: whatever “leaks” due to I/Q mismatch can be cancelled by deliberately “leaking back” the same amount. To compensate for this I/Q leakage, first, the coefficients of the error matrix Aleak should be estimated using off-line or on-line estimation methods. Off-line estimation methods refer to test methods conducted while normal operation of the receiver/transmitter is not in progress, e.g., normal operations have been suspended to inject test signals into the LPF pair and measure the response of those test signals. On-line estimation methods refer to estimation methods which may be conducted during normal receiver/transmitter operation, without suspending the receiver/transmitter normal functions.
Some known off-line estimation methods use sets of one or multiple test frequencies (tones) sent simultaneously into the I/Q chain; measurements are performed by the DSP (FFT) block at the output. Due to the frequency dependency of the LPFs, testing in these known methods is performed over a large number of sets of frequencies (tones) to determine the error matrix Aleak. Other known compensation methods use test-signal based adaptive tuning algorithms, e.g., where testing is repeated iteratively and the Aleak may be gradually adjusted and allowed to converge over time. Examples of known off-line estimation methods are included in the following references:
J. K. Cavers and M. W. Liao, “Adaptive compensation for imbalance and offset losses in direct conversion transceivers,” IEEE Transactions on Vehicular Technology, Vol. 42, No. 4, pp. 581-588 (November 1993);
J. P. F. Glas, “Digital I/Q imbalance compensation in a low-IF receiver,” Proceedings of the IEEE Global Communications Conference, pp. 1461-1466 (1998);
F. E. Churchill, G. W. Ogar, and B. J. Thompson, “The correction of I and Q errors in a coherent processor,” IEEE Transactions on Aerospace and Electronic Systems, Vol. AES-17, pp. 131-137 (January 1981);
K. Pun, J. Franca, and C. Azeredo-Leme, “Wideband digital correction of I and Q mismatch in quadrature radio receivers,” Proceedings of the IEEE International Symposium on Circuits and Systems, pp. V. 661-V. 664 (2000);
X. J. Tao, “Frequency dependent I/Q calibration,” Technical memorandum, Agere Systems, (Oct. 9, 2001); and
L. Der and B. Razavi, “A 2-GHz CMOS image-reject receiver with LMS calibration,” IEEE Journal of Solid-State Circuits, Vol. 38, No. 2, pp. 167-175 (February 2003).
Still other known compensation methods which avoid training (test) signals use blind, on-line adaptive methods to estimate and correct the I/Q imbalance. Examples of known on-line estimation methods are included in the following references:
M. Valkama, M. Renfors, and V. Koivunen, “Advanced methods for I/Q imbalance compensation in communication receivers,” IEEE Transactions on Signal Processing, Vol. 49, No. 10, pp. 2335-2344 (October 2001);
L. Yu and W. M. Snelgrove, “A novel adaptive mismatch cancellation system for quadrature IF radio receivers,” IEEE Transactions on Circuits and Systems-II: Analog and Digital Signal Processing, Vol. 46, No. 6, pp. 789-801 (June 1999); and
K. Pun, J. Franca, and C. Azeredo-Leme, “The correction of frequency-dependent I/Q mismatches in quadrature receivers by adaptive signal separation,” Proceedings of the International Conference on ASIC, pp. 424-427 (2001).
Once Aleak is estimated, a correction matrix Acorr can be found byAcorr=Aleak−1 The correction matrix Acorr may be used to cancel the I/Q leakage. The concept of I/Q leakage compensation is illustrated in FIG. 4 with respect to a single input frequency. Signal Xc 402 (shown in the frequency domain) including no I/Q mismatch (no component at −ω0), is input to an I/Q leakage stage 404. The leakage stage 404 has transfer function Aleak 406. The output of leakage stage 404 is uncorrected signal Yc 408. Signal Yc includes a desirable component at ω0 and an undesirable (or leakage) component at −ω0. Signal Yc 408 is input to a correction stage 410 with transfer function Acorr 412. The output from the correction stage 410 is signal Zc 414. Since Acorr 412 should be tunable and requires high precision, it is usually implemented in the digital domain. The amplitude of the component at −ω0 for compensated signal Zc 414 has been reduced with respect to the corresponding component of the uncompensated signal Yc 408. Due to imperfect error estimation and finite word-length digital correction, some residual I/Q mismatch (component at −ω0) will remain in the corrected output Zc 414.
Note that the above described I/Q imbalance compensation concept is valid for both receiver (RX) and transmitter (TX) applications. The RX implementation applies digital “corrections” or “compensation” following the I/Q leakage in the receiver (Aleak precedes Acorr) . The TX implementation applies digital “pre-distortion” to the digital signal before the signal is subjected to I/Q leakage in the transmitter circuitry (Acorr precedes Aleak)
The I/Q mismatch (δG2(ω) and δP2(ω)) of the filters is frequency dependent, while the I/Q mismatch of the front-end (δG1 and δP1) and the ADCs and/or DACs (δG3) can be considered frequency independent in first order. Therefore, the error matrix Aleak (ω) should be estimated for several frequencies. Thus, the implementation of the correction matrix Acorr (ω) becomes costly since it should be effective over the whole band of frequencies of interest. Known frequency-dependent I/Q estimation/correction methods treat the zero-IF filters as a black box.
Some known methods (previously referenced) to determine the characteristics of this “black box” require the insertion of an extensive range of test signals obtaining measurements over a large number of frequencies. Then elaborate correction models with numerous modeling variables are determined and implemented. The incorporation of elaborate testing circuitry and elaborate correction circuitry (e.g., sometimes including 1,000 to 100,000 or more gates) may consume significant power, may occupy a significant amount of limited circuit board space available, and may add significant cost to the device. In many low cost, limited power, miniature size applications (e.g., hand-held mobile battery operated communication devices) such test signaling and correction circuitry is highly undesirable. In addition, the use of significant amounts of time required to conduct the tests is also highly undesirable. For a battery operated mobile communications device, the time required to conduct the model characterization testing, drains valuable energy from the battery, limiting normal communications operating time between battery recharges. In addition, normal operations are suspended during the off-line test signal measuring intervals possibly disrupting or limiting service. For other communications devices (e.g., a base station, communicating with multiple mobile communications devices) extensive off-line time for testing may be unacceptable, as it may reduce overall system capacity and may interrupt service. Other known methods may use complex adaptive algorithms requiring complex models, numerous adjustment variables, and may require time to converge. Quality of service may be adversely affected during these convergences.
In light of the above discussion, there is a need for better methods and apparatus to provide I/Q compensation, particularly for low pass real filters. Areas which could use improvement include a reduction in complexity of: the leakage estimation test evaluation circuitry, the leakage estimation test method, the compensation model, and the compensation circuitry. New compensation methods and apparatus which reduce the receiver, transmitter, and/or transceiver off-line time for leakage estimation testing would also be beneficial.